This invention relates to an optical information transmission technology, and more particularly, to an optical field transmitter and an optical field transmission system suitable for transmission/reception of an optical multilevel signal transmitted via an optical fiber.
The amount of information that can be transmitted (transmission capacity) via one optical fiber has reached its capacity limit because a wavelength bandwidth of an optical fiber amplifier has been almost used up owing to an increase in number of wavelength channels and a speedup of a modulation speed of an optical signal. In order to further expand the transmission capacity of the optical fiber, it is necessary to improve efficiency of the frequency bandwidth usage by devising a signal modulation method so that a large number of optical signals are packed in a limited frequency bandwidth.
In the world of radio communications, since 1960s, a multilevel modulation technology has realized transmission at such high efficiency that frequency use efficiency exceeds 10. There have conventionally been many studies of multilevel modulation, which is regarded as promising also in the field of optical fiber transmission. For example, R. A. Griffin, et al., “10 Gb/s Optical Differential Quadrature Phase Shift Key (DQPSK) Transmission using GaAs/AlGaAs Integration”, OFC2002, paper PD-FD6, 2002 reports quadrature phase shift keying (QPSK) for performing four-level phase modulation. In addition, N. Kikuchi, K. Mandai, K. Sekine and S. Sasaki, “First experimental demonstration of single-polarization 50-Gbit/s 32-level (QASK and 8-DPSK) incoherent optical multilevel transmission”, in Proc. Optical Fiber Communication Conf. (OFC/NFOEC), Anaheim, Calif., March 2007, PDP21. reports 32-level phase and amplitude modulation that is a combination of four-level amplitude modulation and eight-level phase modulation.
FIGS. 1A to 1D are diagrams describing a complex plane used for the optical transmission and illustrating signal constellations of various known modulation methods. On the complex phase plane (or complex plane, phase plane, IQ plane), there are plotted signal points of various multilevel signals (complex display of an optical field at a decision timing).
FIG. 1A is an explanatory diagram illustrating a signal point on the IQ plane according to a conventional technology.
As illustrated in FIG. 1A, each of the signal points may be displayed by complex Cartesian coordinates (IQ coordinates) or polar coordinates represented by an amplitude r(n) and a phase φ(n) illustrated in FIG. 1A.
FIG. 1B is an explanatory diagram illustrating a signal constellation of quaternary phase shift keying (QPSK) according to a conventional technology.
In FIG. 1B, four ideal signal points (symbols) used for transmitting multilevel signals are displayed on the complex plane. The respective ideal signal points have a fixed amplitude and phase angles φ(n) arranged at four positions 0, π/2, π, and −π/2. When one of the four symbols is transmitted, 2-bit information (00, 01, 11, 10) may be transmitted per symbol. It should be noted that, in a case where this signal is received directly (incoherently) by using optical delay detection, differential quaternary phase shift keying (DQPSK) in which differential precoding is performed in advance is generally employed. However, the signal constellation of QPSK and the signal constellation of DQPSK are the same, and hence QPSK and DQPSK are not distinguished herein.
FIG. 1C is an explanatory diagram illustrating six-level phase modulation in which the phase angles φ(n) are increased to six levels (0, π/3, 2π/3, −π, −2π/3, −π/3) having spacing of π/3 according to a conventional technology.
As illustrated in FIG. 1C, with the six-level phase modulation, information of about 2.58 bits can be transmitted per symbol. However, there have been few examples in which the six-level phase modulation is used in optical communication because of the difficulty of optical delay detection and the mediocre amount of information.
FIG. 1D is an explanatory diagram illustrating sixteen-level quaternary amplitude modulation (16QAM) widely used in radio communications according to a conventional technology.
As illustrated in FIG. 1D, the 16QAM, in which ideal signal points are arranged in lattice, allows four-bit information to be transmitted per symbol. In the example illustrated in FIG. 1D, the Q-axis coordinate represents a value of upper two bits (10xx, 11xx, 01xx, 00xx), and the I-axis coordinate represents a value of lower two bits (xx10, xx11, xx01, xx00). It is known that in this signal constellation, a distance between the signal points increases to improve the receiver sensitivity. It is reported that in optical communications, the quadrature amplitude modulation similar to 16QAM can be realized by using a coherent optical receiver. For example, J. Hongou, K. Kasai, M. Yoshida and M. Nakazawa, “1 Gsymbol/s, 64 QAM Coherent Optical Transmission over 150 km with a Spectral Efficiency of 3 Bit/s/Hz”, in Proc. Optical Fiber Communication Conf. (OFC/NFOFEC), Anaheim, Calif., March 2007, paper OMP3. reports an experimental example of transmission/reception of a 64QAM signal. The coherent receiver employs a format that uses a local light source disposed within the receiver in order to detect the phase angle of the optical signal.
Here, description is made of a coherent reception format which is one of conventional technologies for an optical multilevel receiver, for example, a coherent optical field receiver reported in M. G. Taylor, “Coherent detection method using DSP to demodulate signal and for subsequent equalization of propagation impairments”, paper We4. P. 111, ECOC 2003, 2003.
FIG. 2 is a block diagram illustrating a coherent optical field receiver of a polarization diversity type, which simultaneously receives information on two polarizations of the optical signal according to a conventional technology.
An optical multilevel signal transmitted through an optical fiber transmission line is amplified by an optical amplifier 117, and then input to a polarization beam splitter 102-1 as an input optical signal 101. The input optical signal 101 is split into a horizontal (S) polarization component 105 and a vertical (P) polarization component 106, which are input to coherent field detector front ends 100-1 and 100-2, respectively.
In the coherent field detector front end 100-1, a local laser source 103 which emits an optical signal having a wavelength substantially the same as the input optical signal 101 is used as a reference of an optical phase. Local light 104-1 output from the local laser source 103 is split by a polarization beam splitter 102-2 into two beams of local light 104-2 and local light 104-3, which are input to the coherent field detector front ends 100-1 and 100-2, respectively.
Inside the coherent field detector front end 100-1, an optical phase diversity circuit 107 combines the S polarization component 105 of the input optical signal and the local light 104-2 to generate an inphase (I) component output light 108 including an inphase component of the local light and the optical multilevel signal, and a quadrature (Q) component output light 109 including a quadrature component of the local light and the optical multilevel signal. The inphase (I) component output light 108 and the quadrature (Q) component output light 109 are received by balanced optical detectors 110-1 and 110-2, respectively, to be converted into electric signals, which are then time-sampled by A/D converters 111-1 and 111-2 to become digitized output signals 112-1 and 112-2, respectively.
In the following description, as illustrated in FIG. 1A, the optical field of the input optical signal 101 is represented as r(n)exp(jφ(n)), and the optical field of the local light 104-2 and the local light 104-3 is assumed to be 1 (originally, an optical frequency component is included, but the optical frequency component is omitted). Here, “r” represents an amplitude of the optical field, “φ” represents a phase of the optical field, and “n” represents a sampling number. The local light 104-2 and the local light 104-3 actually have random phase noise and a slight difference frequency component with respect to signal light. However, the phase noise and the difference frequency component exhibit temporally slow phase rotation, and may be eliminated by digital signal processing. Therefore, the phase noise and the difference frequency component are ignored.
The balanced optical detectors 110-1 and 110-2 each perform homodyne detection on the input optical signal 101 with the local light 104-2, and output an inphase component and a quadrature component of the optical field of the optical multilevel signal 101 by taking the local light 104-2 and the local light 104-3 as a reference, respectively. Therefore, the output signal 112-1 from the A/D converter 111-1 is I(n)=r(n)cos(φ(n)), and the output signal 112-2 from the A/D converter 111-2 is Q(n)=r(n)sin(φ(n)). However, for simplification, constants including a conversion factor are all set to “1”.
As described above, the coherent optical field receiver can easily obtain all information pieces indicating the optical field r(n)exp(φ(n)) (both I component and Q component) from the input optical signal 101, thereby allowing the reception of a complex modulated optical multilevel signal.
A digital signal processing circuit 113, which is a complex field signal processing circuit, gives an inverse function to a linear degradation (for example, chromatic dispersion) or the like exerted upon the optical signal during transmission, to thereby enable cancellation of influences of the linear degradation. Further, processings such as retiming and resampling are performed to output a demodulated received field 116-1.
The coherent field detector front end 100-1 can obtain field information on the S polarization component of the input optical signal 101 as described above. However, a polarization state of the transmitted optical signal changes at random during the optical fiber transmission, and hence a part or all of the transmitted light may be converted to the orthogonal P polarization, which leads to a fear that the coherent field detector front end 100-1 cannot receive the field information on the S polarization component. To avoid this problem, when the coherent optical field receiver is used, polarization diversity reception is used, in which the S polarization and the P polarization of the received light are received by different receivers and recombined. Specifically, the other coherent field detector front end 100-2 is used to receive the P polarization component of the input optical signal 101 to obtain AD-converted output signals 112-3 and 112-4. The digital signal processing circuit 113 resolves the change of the polarization state by subjecting the output signals 112-1 to 112-4 (that is, the I components and the Q components of the polarizations) to equalization processing such as conversion of the polarization state and polarization mode dispersion, to thereby obtain the demodulated received field 116-1.
Subsequently, a symbol decision circuit 114, which uses Euclidean distances, compares the received signal constellation with the ideal signal constellation illustrated in FIG. 1B, for example, and decides which ideal signal point has been received, to thereby output a multilevel symbol string 115.
In coherent reception, it is generally known that a noise distribution of the received signal is isotropic on a signal plane. This is a state in which, as illustrated in FIG. 1B, a noise distribution is represented by a circle (hatched portion) centered on each signal point. In such case, decision based on the Euclidean distances is used for decision of the received signal, to thereby enable reception at the highest sensitivity.
FIG. 3A to FIG. 3C are each an explanatory diagram of a signal constellation of a conventional optical multilevel modulation method and a decision area for a received symbol based on Euclidean distances.
FIG. 3A is an explanatory diagram illustrating a signal constellation and a decision area for a received symbol based on Euclidean distances of the four-level phase modulation of the conventional technology.
When the received signal is the quaternary phase shift keying (QPSK), as illustrated in FIG. 3A, Euclidean distances d(X,A), d(X,B), d(X,C), and d(X,D) between the received field X and four ideal signal points A to D are calculated on the complex plane, and the ideal signal point (C in FIG. 3A) having the smallest Euclidean distance is decided as the received signal point. It should be noted that the Euclidean distance is a length of a line connecting two points in the figure. Meanwhile, the bold lines in the figure are boundary lines each at an equal distance from two adjacent signal points and serve as boundaries of the decision area for the signal point. For example, when the received field X falls in the area indicated by the vertical lines (phase angle: 3π/4 to −3π/4), the received symbol is decided as C.
FIG. 3B is an explanatory diagram illustrating a signal constellation, a decision area, and boundary lines of the six-level phase modulation of the conventional technology.
FIG. 3C is an explanatory diagram illustrating a signal constellation, a decision area, and boundary lines of the sixteen-level phase modulation of the conventional technology.
As described above, the decision using the Euclidean distances has a feature that a decision area for each symbol is formed by lines each dividing an area between two signal points into exact halves.
It should be noted that, with respect to the coherent optical field receiver of the polarization diversity type of FIG. 2, an example in which two receivers are used to extract information on a transmission signal of one polarization has been described. However, employment of a polarization multiplexing format is also considered, in which mutually independent pieces of information are multiplexed with two orthogonal polarizations to be transmitted as transmission signals. In the polarization multiplexing, two transmitters for the X polarization and the Y polarization are provided on the transmit side so that both the X polarization and the Y polarization are subjected to the polarization multiplexing to be transmitted long-distance through the optical fiber transmission line, and both the X polarization and the Y polarization are received at the same time by the coherent optical field receiver of the polarization diversity type illustrated in FIG. 2. The digital signal processing circuit 113 performs orthogonal separation of the polarization components and equalization processing for the polarization mode dispersion, and extracts the demodulated received field 116-1 of the original X polarization component and a demodulated received field 116-2 of the original Y polarization component separately. The symbol decision circuit 114, which uses the Euclidean distances, performs symbol decision for each of the components, and demodulates two sets of multilevel symbol strings 115.
FIG. 4 is a block diagram illustrating a configuration of a phase pre-integration incoherent optical multilevel transmission system according to a conventional technology.
The phase pre-integration incoherent optical multilevel transmission system illustrated in FIG. 4 easily realizes transmission of the optical multilevel signal on the complex plane using optical delay detection with no local light source.
An unmodulated laser beam output from a laser source 210 is input to an optical field modulator 211 within a phase pre-integration optical field transmitter 200, and a transmission optical multilevel signal 213 subjected to required field modulation is output from an output optical fiber 212. A binary digital information signal 201 to be transmitted is converted into a complex multilevel information signal 203 within a multilevel encoder 202. The complex multilevel information signal 203 is a digital electric multilevel signal represented as (i, q) on a two-dimensional IQ plane, and a real part i and an imaginary part q of the signal are output at every time interval T (=symbol time). In an explanatory diagram illustrated in FIG. 4, a 16QAM signal illustrated in the balloon is used as an example of the complex multilevel information signal 203.
The complex multilevel information signal 203 is input to a phase pre-integration unit 204, in which only phase components of the input signal are digitally integrated at time intervals T and converted into a phase pre-integration complex multilevel information signal 205. Here, when Ei(n)=(i(n),q(n)) indicating the input complex multilevel information signal 203 is converted into polar coordinates on the complex plane, the signal can be represented as, for example, Ei(n)=i(n)+jq(n)=r(n)exp(jφ(n)) (j is an imaginary part unit). In this expression, n is a symbol number of the digital signal, r(n) is a symbol amplitude of the digital signal, and φ(n) is a phase angle. The phase pre-integrated signal to be output at this time can also be represented in polar coordinates as Eo(n)=i′(n)+jq′(n)=r(n)exp(jθ(n))=r(n)exp(jΣφ(n)). In this expression, θ(n) is a phase angle of the output signal, and Σφ(n) is a value obtained by accumulating past phase angles φ(1) to φ(n) at every time T. The output phase pre-integrated signal is again converted into a Cartesian coordinate system, and then output as the phase pre-integration complex multilevel information signal 205. Inside the balloon, the phase pre-integration complex multilevel information signal 205 is represented on the complex plane and has a concentric signal constellation that is significantly different from the complex multilevel information signal 203, which is the original 16QAM signal, after phase pre-integration operation.
The phase pre-integration complex multilevel information signal 205 is input to a sampling speed conversion circuit 206 and complemented so that the sampling speed becomes 2 samples/symbol or more. Thereafter, an inverse function of a degradation developed in an optical transmission line 214 or the like is applied to the phase pre-integration complex multilevel information signal 205 by a preequalization circuit 207, and then divided into a real part i″ and an imaginary part q″, which are converted into high-speed analog signals by DA converters 208-1 and 208-2, respectively. Those two high-speed analog signals are amplified by driver circuits 209-1 and 209-2, and then input to two modulation terminals I and Q of the optical field modulator 211. As a result, the transmission optical multilevel signal 213 can be generated with the preequalization phase integrated signals (i″(n), q″(n)) in the inphase component I and the quadrature component Q of the optical field. It should be noted that the optical field of the transmission optical multilevel signal 213 is (i″(n)+jq″(n))exp(jω(n)), and ω(n) is an optical angular frequency of the laser source 210. That is, the transmission optical multilevel signal 213 is (i″(n), q″(n)) when represented in the equalization low band where the optical frequency component is removed.
The transmission optical multilevel signal 213 is transmitted through the optical fiber transmission line 214, subjected to a transmission degradation by chromatic dispersion or the like of the optical fiber and amplification by the optical amplifier 117, and thereafter input to an incoherent optical field receiver 220 as a received optical multilevel signal 215. The transmission degradation is mutually canceled by the inverse function applied by the preequalization circuit 207 in advance, and therefore the optical field of the received signal is equal to the phase pre-integration complex multilevel information signal 205.
The received optical multilevel signal 215 is split into three optical signal paths by an optical splitter 222 in an incoherent field detector front end 221 to be input to a first optical delay detector 223-1, a second optical delay detector 223-2, and an optical intensity detector 225. The first optical delay detector 223-1 is set so that one of two optical paths has a delay time Td that is substantially equal to a symbol time T of the received optical multilevel information signal, and so that a difference of optical phase between those optical paths becomes 0. Further, the second optical delay detector 223-2 is set so that one of two optical paths has a delay time Td=T, and so that a difference of optical phase between those optical paths becomes π/2. Two output light beams of the first optical delay detector 223-1 and the second optical delay detector 223-2 are converted into electric signals by balanced optical detectors 224-1 and 224-2, respectively, and further into digital signals dI(n) and dQ(n) by A/D converters 226-1 and 226-2, respectively. Further, an output electric signal from the optical intensity detector 225 is also converted into a digital signal P(n) by an AD converter 226-3.
Thereafter, the digital signals dI(n) and dQ(n) are input to an inverse tangent circuit 227. The inverse tangent circuit 227 conducts inverse tangent operation of two arguments with dI(n) as an X component and dQ(n) as a Y component, and calculates a phase angle of the digital signals dI(n) and dQ(n). When the optical field of the received multilevel signal 215 is described as r(n)exp(jθ(n)), the digital signals dI(n) and dQ(n) can be represented as follows:
dI(n)∝r(n)r(n−1)cos(Δθ(n)); and
dQ(n)∝r(n)r(n−1)sin(Δθ(n)),
based on the principle of the optical delay detection. In those expressions, Δθ(n) is a phase difference (θ(n)−θ(n−1)) from a symbol immediately before a received n-th optical field symbol. dI(n) and dQ(n) are a sine component and a cosine component of Δθ(n), respectively, and hence the inverse tangent circuit 227 conducts 4-quadrant inverse tangent (arc tan) operation so as to calculate Δθ(n).
It should be noted that, in this configuration, because the phase pre-integration is conducted at the transmit side as described above, a phase angle of the received optical field signal is θ(n)=Σφ(n). Hence, an output signal of the inverse tangent circuit 227 is Δθ(n)=Σφ(n)−Σφ(n−1)=φ(n), and a phase component φ(t) of the original complex multilevel information signal 203 can be extracted.
On the other hand, the output signal P(n) of the optical intensity detector 225 is input to a square root circuit 228 so as to obtain an original field amplitude r(n)=sqrt(P(n)) as an output. Therefore, when the obtained amplitude component r(n) and phase component φ(n) are input to a Cartesian coordinate converter circuit 229, a Cartesian coordinate representation (i, q)=r(n)exp(jφ(n)) is obtained as the demodulated received field 116. This is the same signal constellation as that of the original complex multilevel information signal 203. Therefore, when the obtained signal is input to the symbol decision circuit 114, which uses the Euclidean distances, to perform the symbol decision, the multilevel symbol string 115 can be generated again.
It should be noted that, in the phase pre-integration transmission format, the signal constellation of the optical field transmitted by the transmitter (which is the same as the signal constellation of the phase pre-integration complex multilevel information signal 205) and the signal constellation of the demodulated received field 116 in the receiver are different as described above. Hereinafter, this invention mainly focuses on a decision format of the received field. Therefore, the terms “the signal constellation” and “decision” as used herein regarding the phase pre-integration format refer to the signal constellation of the demodulated received field 116 (or the complex multilevel information signal 203).    Non Patent Literature 1: R. A. Griffin, et al., “10 Gb/s Optical Differential Quadrature Phase Shift Key (DQPSK) Transmission using GaAs/AlGaAs Integration”, OFC2002, paper PD-FD6, 2002    Non Patent Literature 2: N. Kikuchi, K. Mandai, K. Sekine and S. Sasaki, “First experimental demonstration of single-polarization 50-Gbit/s 32-level (QASK and 8-DPSK) incoherent optical multilevel transmission”, in Proc. Optical Fiber Communication Conf. (OFC/NFOEC), Anaheim, Calif., March 2007, PDP21    Non Patent Literature 3: J. Hongou, K. Kasai, M. Yoshida and M. Nakazawa, “1 Gsymbol/s, 64 QAM Coherent Optical Transmission over 150 km with a Spectral Efficiency of 3 Bit/s/Hz”, in Proc. Optical Fiber Communication Conf. (OFC/NFOFEC), Anaheim, Calif., March 2007, paper OMP3    Non Patent Literature 4: M. G. Taylor, “Coherent detection method using DSP to demodulate signal and for subsequent equalization of propagation impairments”, paper We4. P. 111, ECOC 2003, 2003